A switched reluctance motor is an example of an electrical load in which the inductance is not constant with time or machine operation, i.e., rotor angle. FIG. 1(a) illustrates a typical three-phase switched reluctance (SR) machine and FIGS. 1(b) and (c) illustrate well-known examples of electronic switching circuits which may be used to control an SR machine. The SR machine essentially consists of a stator s defining stator poles 1, 1', 2, 2', 3, 3' on which are wound phase windings w, of which only one is shown in association with a set of poles 2, 2'. The machine also has a rotor with salient poles 4, 4' and 5, 5'. The electronic circuits are arranged to supply unidirectional currents to the phase windings w.
In the control circuits of FIGS. 1(b) and (c), each phase winding of the machine in FIG. 1(a) is associated with a circuit leg comprising at least one electronic switch t in series with each winding across a dc supply Vs. A general treatment of the principles of SR machines is given in the paper `The Characteristics, Design and Applications of Switched Reluctance Motors and Drives` by Stephenson et al. presented at PCIM '93 Conference and Exhibition at Nurnberg, Germany, Jun. 21st-24th 1993.
Proposed means for controlling an SR machine to run at different speeds include operation in three characteristic regions which can be designated as `low-speed`, `medium-speed` and `high-speed` regions. For background and explanation on this see `Variable-Speed Switched Reluctance Motors` by Lawrenson et al. IEE Proceedings Part B, Vol. 127, No. 4, Jul. 1980. In the low-speed region the current is controlled by the well-known method of `chopping`. The present invention relates to improvements in the means of implementing chopping control of the phase current. FIGS. 2(a), (b) and (c) represent one phase leg of the circuit shown in FIG. 1(b) redrawn for clarity with the phase leg in three possible states, here referred to as `ON`, `FW` (`freewheeling`) and `OFF`. The circuit is in the ON state when both power switches are closed and the full dc bus voltage Vs is applied to the phase winding w, increasing the magnetic flux F at the maximum possible rate (see FIG. 2(a)).
When flux has been established in the winding in this way and either one of the switches is open, the current is said to freewheel (see FIG. 2(b)), i.e. the circuit is in the FW state with only one switch closed, the current flowing through this switch and one diode. The effective winding voltage under these conditions is determined by the small voltage across the switch, the diode and the winding resistance. The flux F thus falls relatively slowly, as shown in FIG. 2(b).
The circuit is in the OFF condition when both switches are open and the phase current is carried by the diodes. The winding then has the full dc bus voltage applied in reverse so that the flux F will fall until the current is zero and the diodes become non-conducting (see FIG. 2(c)).
The simplest method of current chopping is to alternate between ON and OFF states to maintain the mean current level near a desired value. This is shown in FIG. 3(a). Chopping between the ON and OFF states is acceptable at low power levels where the switches (e.g. semiconductor switches such as metal oxide silicon dioxide field effect transistors or insulated gate bipolar transistors) can switch at ultrasonic frequencies. This is advantageous in terms of limiting acoustic noise.
At higher power levels however, the losses (both in the semi-conductor switches and other components) associated with ultrasonic switching become large and it is usually necessary to reduce the switching frequency. If the ON/OFF strategy were used the current (and the flux) excursions at these reduced frequencies might be large, resulting in a rise in objectionable acoustic noise and, possibly, control problems as well. For these reasons the FW state is often incorporated into the switching pattern, enabling retention of relatively small current excursions even though the switching frequency is reduced. This is shown in FIG. 3(b) for a `motoring` mode and in FIG. 3(c) for a `generating` mode.
The behaviour of the winding current in a reluctance motor during freewheel (FW) is determined not only by the applied voltage but also by the variation of the phase inductance which is a function of the rotation of the machine. Inductance is defined as flux linkage per unit current so that L=F/I, hence I=F/L. If the effective winding voltage during freewheeling is small, the flux can be considered constant over a short period and the current during that period will therefore follow the reciprocal of the inductance profile, i.e. the freewheeling current will reduce when the load inductance rises and increase when it falls. At low rotational speeds, however, the rate of change of inductance with time (dL/dt) is small and the reduction in flux with time during freewheeling becomes significant over a machine phase period. At very low speeds, the freewheeling current will fall even when the inductance is decreasing, because a small winding voltage causes the flux to fall faster than the inductance. The behaviour of the freewheeling current under these differing conditions has important consequences for the current control system, as the controller has to be able to function correctly in all these different conditions. In a typical control system for a switched reluctance motor, the phase windings W will be energised whilst their inductance is increasing with respect to rotor angle (the motoring mode) or when the inductance is falling with respect to rotor angle (the braking or generating mode).
In the prior art, the control considerations described above have been implemented in a variety of ways, each with its own advantages and disadvantages.
One of the simplest methods is to use one comparator for each phase, the current feedback being compared directly with the reference (demand) value. A hysteresis band separates the switching points. The width of this band is usually varied to provide a suitable compromise between current excursions, switching frequency and acoustic noise. A typical circuit is illustrated in FIG. 4.
In a motoring mode (i.e. in which the net power flow is from the source to the load), the controller regulates current by alternating between the ON and FW states, with both switches switched off at the end of the phase period. FIG. 5(a) shows a typical current waveform. The change from motoring to a generating mode requires the MOT/GEN logic signal shown in FIG. 4. In the generating mode (i.e. in which the net power flow is from the load to the source), the controller chops by alternating between FW and OFF, though the ON state must first be used to build up flux (and hence phase current) to the required working value. FIG. 5(b) shows a typical current waveform.
While this system has the advantages of using only one comparator and only one hysteresis band, it has a number of disadvantages. These are particularly apparent at low-speeds whilst generating. Because the generating freewheel current rises only if the speed is high enough, at low speeds (particularly at high currents when the effective freewheel winding voltage is greatest) the current decays below the required value, as illustrated in FIG. 5(c). The output of the generator is then reduced. A further drawback is the dependence on a logic signal for switching between motoring and generating modes. This signal may be difficult to generate reliably, especially during transient conditions. This may lead to loss of control of the current, resulting in nuisance tripping or even switch failure.
A second proposed system seeks to overcome these difficulties by using two comparators. The two comparators have the same reference, but the hysteresis band of one spans that of the other, as shown in FIG. 6(a). Essentially, the `outer` comparator is used to modify the behaviour of the circuitry controlled by the `inner` comparator so that, for most of the time, chopping occurs between the switching points of the inner comparator only. The power switches are switched off at the end of the phase period as before. In the motoring mode, this system behaves like the single comparator case and chopping is controlled by the inner hysteresis band only, as shown in FIG. 6(b).
In the generating mode, the controller starts in the same mode with both switches closed until the current reaches the upper level of the inner comparator, whereupon one switch is opened and the phase freewheels. The freewheel current rises further (because the system is generating) until the upper level of the outer comparator is reached whereupon the second switch is also opened. The inner comparator now operates in the generating mode, selecting either FW or OFF. Chopping between FW and OFF continues on the inner band unless the current fails to rise while the circuit is in the FW state. If the current falls to the lower level of the outer comparator, both switches are closed, and the resulting ON state raises the current into the control band of the inner comparator. This is illustrated in FIG. 6(c).
This system has some advantages. For example, it keeps control of the current at all times, irrespective of whether the drive is motoring or generating and avoids the problems of decaying currents at low speeds associated with the single comparator solution.
However, it has disadvantages which may render it of little value in some applications. It suffers from relatively large transient excursions of current to the outer hysteresis bands which may be objectionable in some applications. This is made worse by the fact that the outer band must generally be wide so as to keep it reliably distinct from the inner one over the full working range of current despite noise, drift or other possible sources of signal corruption.